Apparatus for transmitting and receiving a signal and method of transmitting and receiving a signal

ABSTRACT

The present invention relates to a method of transmitting and receiving signals and a corresponding apparatus. One aspect of the present invention relates to a method of receiving a signal, which includes interleaving in an appropriate manner for a channel bonding system. The interleaving can allow decoding a user requested service at a random tuner window position.

TECHNICAL FIELD

The present invention relates to a method for transmitting and receivinga signal and an apparatus for transmitting and receiving a signal, andmore particularly, to a method for transmitting and receiving a signaland an apparatus for transmitting and receiving a signal, which arecapable of improving data transmission efficiency.

BACKGROUND ART

As a digital broadcasting technology has been developed, users havereceived a high definition (HD) moving image. With continuousdevelopment of a compression algorithm and high performance of hardware,a better environment will be provided to the users in the future. Adigital television (DTV) system can receive a digital broadcastingsignal and provide a variety of supplementary services to users as wellas a video signal and an audio signal.

Digital Video Broadcasting (DVB)-C2 is the third specification to joinDVB's family of second generation transmission systems. Developed in1994, today DVB-C is deployed in more than 50 million cable tunersworldwide. In line with the other DVB second generation systems, DVB-C2uses a combination of Low-density parity-check (LDPC) and BCH codes.This powerful Forward Error correction (FEC) provides about 5 dBimprovement of carrier-to-noise ratio over DVB-C. Appropriatebit-interleaving schemes optimize the overall robustness of the FECsystem. Extended by a header, these frames are called Physical LayerPipes (PLP). One or more of these PLPs are multiplexed into a dataslice. Two dimensional interleaving (in the time and frequency domains)is applied to each slice enabling the receiver to eliminate the impactof burst impairments and frequency selective interference such as singlefrequency ingress.

DISCLOSURE OF INVENTION Technical Problem

With the development of these digital broadcasting technologies, arequirement for a service such as a video signal and an audio signalincreased and the size of data desired by users or the number ofbroadcasting channels gradually increased.

Technical Solution

Accordingly, the present invention is directed to a method fortransmitting and receiving a signal and an apparatus for transmittingand receiving a signal that substantially obviate one or more problemsdue to limitations and disadvantages of the related art.

An object of the present invention is to provide a method oftransmitting broadcasting signal to a receiver, comprising: mappingpreamble data bits into preamble data symbols and data bits into datasymbols; building at least one data slice based on the data symbols;time-interleaving the data symbols at a level of the data slice;building a signal frame based on the preamble data symbols and the dataslice, the preamble data symbols comprising Layer 1(L1) signalinginformation for signaling the data slice; modulating the built signalframe by an Orthogonal Frequency Division Multiplexing (OFDM) method;and transmitting the modulated signal frame.

Another aspect of the present invention provides a method of receivingbroadcasting signal, comprising; demodulating the received signal by useof an Orthogonal Frequency Division Multiplexing(OFDM) method; obtaininga signal frame from the demodulated signals, the signal frame comprisingpreamble symbols and data symbols, wherein the preamble symbols haveLayer 1(L1) signaling information, wherein the data symbols are dividedinto at least one data slices; frequency-deinterleaving the data symbolsat a level of the data slice; demapping the time-deinterleaved datasymbols into bits; and decoding the bits by LDPC(low density paritycheck) decoding scheme.

Yet another aspect of the present invention provides a transmitter oftransmitting broadcasting signal to a receiver, the transmittercomprising: a mapper configured to map preamble data bits into preambledata symbols and data bits into data symbols; a data slice builderconfigured to build at least one data slice based on the data symbols; atime-interleaver configured to time-interleave the data symbols at alevel of the data slice; a frame builder configured to build a signalframe based on the preamble data symbols and the data slice, thepreamble data symbols comprising Layer 1(L1) signaling information forsignaling the data slice; a modulator configured to Modulate the builtsignal frame by an Orthogonal Frequency Division Multiplexing (OFDM)method; and a transmission unit configured to transmit the modulatedsignal frame.

Yet another aspect of the present invention provides a receiver ofreceiving broadcasting signal, the receiver comprising: a demodulatorconfigured to demodulate the received signal by use of an OrthogonalFrequency Division Multiplexing(OFDM) method; a frame Parser configuredto obtain a signal frame from the demodulated signals, the signal framecomprising preamble symbols and data symbols, wherein the preamblesymbols have Layer 1(L1) signaling information, wherein the data symbolsare divided into at least one data slice; a frequency-deinterleaverconfigured to frequency-deinterleave the data symbols at a level of thedata slice; a demapper configured to demap the time-deinterleaved datasymbols into bits; and a decoder configured to decode the bits byLDPC(Low Density Parity Check) decoding scheme.

BRIEF DESCRIPTION OF DRAWINGS

The accompanying drawings, which are included to provide a furtherunderstanding of the invention and are incorporated in and constitute apart of this application, illustrate embodiment(s) of the invention andtogether with the description serve to explain the principle of theinvention. In the drawings:

FIG. 1 is an example of 64-Quadrature amplitude modulation (QAM) used inEuropean DVB-T.

FIG. 2 is a method of Binary Reflected Gray Code (BRGC).

FIG. 3 is an output close to Gaussian by modifying 64-QAM used in DVB-T.

FIG. 4 is Hamming distance between Reflected pair in BRGC.

FIG. 5 is characteristics in QAM where Reflected pair exists for each Iaxis and Q axis.

FIG. 6 is a method of modifying QAM using Reflected pair of BRGC.

FIG. 7 is an example of modified 64/256/1024/4096-QAM.

FIGS. 8-9 are an example of modified 64-QAM using Reflected Pair ofBRGC.

FIGS. 10-11 are an example of modified 256-QAM using Reflected Pair ofBRGC.

FIGS. 12-13 are an example of modified 1024-QAM using Reflected Pair ofBRGC(0˜511).

FIGS. 14-15 are an example of modified 1024-QAM using Reflected Pair ofBRGC(512˜1023).

FIGS. 16-17 are an example of modified 4096-QAM using Reflected Pair ofBRGC(0˜511).

FIGS. 18-19 are an example of modified 4096-QAM using Reflected Pair ofBRGC(512˜1023).

FIGS. 20-21 are an example of modified 4096-QAM using Reflected Pair ofBRGC(1024˜1535).

FIGS. 22-23 are an example of modified 4096-QAM using Reflected Pair ofBRGC(1536˜2047).

FIGS. 24-25 are an example of modified 4096-QAM using Reflected Pair ofBRGC(2048˜2559).

FIGS. 26-27 are an example of modified 4096-QAM using Reflected Pair ofBRGC(2560˜3071).

FIGS. 28-29 are an example of modified 4096-QAM using Reflected Pair ofBRGC(3072˜3583).

FIGS. 30-31 are an example of modified 4096-QAM using Reflected Pair ofBRGC(3584˜4095).

FIG. 32 is an example of Bit mapping of Modified-QAM where 256-QAM ismodified using BRGC.

FIG. 33 is an example of transformation of MQAM into Non-uniformconstellation.

FIG. 34 is an example of digital transmission system.

FIG. 35 is an example of an input processor.

FIG. 36 is an information that can be included in Base band (BB).

FIG. 37 is an example of BICM.

FIG. 38 is an example of shortened/punctured encoder.

FIG. 39 is an example of applying various constellations.

FIG. 40 is another example of cases where compatibility betweenconventional systems is considered.

FIG. 41 is a frame structure which comprises preamble for L1 signalingand data symbol for PLP data.

FIG. 42 is an example of frame builder.

FIG. 43 is an example of pilot insert (404) shown in FIG. 4.

FIG. 44 is a structure of SP.

FIG. 45 is a new SP structure or Pilot Pattern (PP) 5.

FIG. 46 is a suggested PP5′ structure.

FIG. 47 is a relationship between data symbol and preamble.

FIG. 48 is another relationship between data symbol and preamble.

FIG. 49 is an example of cable channel delay profile.

FIG. 50 is scattered pilot structure that uses z=56 and z=112.

FIG. 51 is an example of modulator based on OFDM.

FIG. 52 is an example of preamble structure.

FIG. 53 is an example of Preamble decoding.

FIG. 54 is a process for designing more optimized preamble.

FIG. 55 is another example of preamble structure

FIG. 56 is another example of Preamble decoding.

FIG. 57 is an example of Preamble structure.

FIG. 58 is an example of L1 decoding.

FIG. 59 is an example of analog processor.

FIG. 60 is an example of digital receiver system.

FIG. 61 is an example of analog processor used at receiver.

FIG. 62 is an example of demodulator.

FIG. 63 is an example of frame parser.

FIG. 64 is an example of BICM demodulator.

FIG. 65 is an example of LDPC decoding using shortening/puncturing.

FIG. 66 is an example of output processor.

FIG. 67 is an example of L1 block repetition rate of 8 MHz.

FIG. 68 is an example of L1 block repetition rate of 8 MHz.

FIG. 69 is a new L1 block repetition rate of 7.61 MHz.

FIG. 70 is an example of L1 signaling which is transmitted in frameheader.

FIG. 71 is preamble and L1 Structure simulation result.

FIG. 72 is an example of symbol interleaver.

FIG. 73 is an example of an L1 block transmission.

FIG. 74 is another example of L1 signaling transmitted within a frameheader.

FIG. 75 is an example of frequency or time interleaving/deinterleaving.

BEST MODE FOR CARRYING OUT THE INVENTION

Reference will now be made in detail to the preferred embodiments of thepresent invention, examples of which are illustrated in the accompanyingdrawings. Wherever possible, the same reference numbers will be usedthroughout the drawings to refer to the same or like parts.

In the following description, the term “service” is indicative of eitherbroadcast contents which can be transmitted/received by the signaltransmission/reception apparatus.

Quadrature amplitude modulation (QAM) using Binary Reflected Gray Code(BRGC) is used as modulation in a broadcasting transmission environmentwhere conventional Bit Interleaved Coded Modulation (BICM) is used. FIG.1 shows an example if 64-QAM used in European DVB-T.

BRGC can be made using the method shown in FIG. 2. An n bit BRGC can bemade by adding a reverse code of (n-1) bit BRGC (i.e., reflected code)to a back of (n-1) bit, by adding 0s to a front of original (n-1) bitBRGC, and by adding is to a front of reflected code. The BRGC code madeby this method has a Hamming distance between adjacent codes of one (1).In addition, when BRGC is applied to QAM, the Hamming distance between apoint and the four points which are most closely adjacent to the point,is one (1) and the Hamming distance between the point and another fourpoints which are second most closely adjacent to the point, is two (2).Such characteristics of Hamming distances between a specificconstellation point and other adjacent points can be dubbed as Graymapping rule in QAM.

To make a system robust against Additive White Gaussian Noise (AWGN),distribution of signals transmitted from a transmitter can be made closeto Gaussian distribution. To be able to do that, locations of points inconstellation can be modified. FIG. 3 shows an output close to Gaussianby modifying 64-QAM used in DVB-T. Such constellation can be dubbed asNon-uniform QAM (NU-QAM).

To make a constellation of Non-uniform QAM, Gaussian CumulativeDistribution Function (CDF) can be used. In case of 64, 256, or 1024QAM, i.e., 2̂N AMs, QAM can be divided into two independent N-PAM. Bydividing Gaussian CDF into N sections of identical probability and byallowing a signal point in each section to represent the section, aconstellation having Gaussian distribution can be made. In other words,coordinate xj of newly defined non-uniform N-PAM can be defined asfollows:

$\begin{matrix}{{{\int_{- \infty}^{X_{j}}{\frac{1}{\sqrt{2\pi}}^{- \frac{X^{2}}{2}}\ {x}}} = p_{j}},{P_{j} \in \left\{ {\frac{1}{2N},\frac{3}{2N},\ldots \mspace{14mu},\frac{{2N} - 1}{2N}} \right\}}} & \left( {{Eq}.\mspace{14mu} 1} \right)\end{matrix}$

FIG. 3 is an example of transforming 64 QAM of DVB-T into NU-64 QAMusing the above methods. FIG. 3 represents a result of modifyingcoordinates of each I axis and Q axis using the above methods andmapping the previous constellation points to newly defined coordinates.In case of 32, 128, or 512 QAM, i.e., cross QAM, which is not 2̂N QAM, bymodifying Pj appropriately, a new coordinate can be found.

One embodiment of the present invention can modify QAM using BRGC byusing characteristics of BRGC. As shown in FIG. 4, the Hamming distancebetween Reflected pair in BRGC is one because it differs only in one bitwhich is added to the front of each code. FIG. 5 shows thecharacteristics in QAM where Reflected pair exists for each I axis and Qaxis. In this figure, Reflected pair exists on each side of the dottedblack line.

By using Reflected pairs existing in QAM, an average power of a QAMconstellation can be lowered while keeping Gray mapping rule in QAM. Inother words, in a constellation where an average power is normalized as1, the minimum Euclidean distance in the constellation can be increased.When this modified QAM is applied to broadcasting or communicationsystems, it is possible to implement either a more noise-robust systemusing the same energy as a conventional system or a system with the sameperformance as a conventional system but which uses less energy.

FIG. 6 shows a method of modifying QAM using Reflected pair of BRGC.FIG. 6 a shows a constellation and FIG. 6 b shows a flowchart formodifying QAM using Reflected pair of BRGC. First, a target point whichhas the highest power among constellation points needs to be found.Candidate points are points where that target point can move and are theclosest neighbor points of the target point's reflected pair. Then, anempty point (i.e., a point which is not yet taken by other points)having the smallest power needs to be found among the candidate pointsand the power of the target point and the power of a candidate point arecompared. If the power of the candidate point is smaller, the targetpoint moves to the candidate point. These processes are repeated untilan average power of the points on constellation reaches a minimum whilekeeping Gray mapping rule.

FIG. 7 shows an example of modified 64/256/1024/4096-QAM. The Graymapped values correspond to FIGS. 8˜31 respectively. In addition tothese examples, other types of modified QAM which enables identicalpower optimization can be realized. This is because a target point canmove to multiple candidate points. The suggested modified QAM can beapplied to, not only the 64/256/1024/4096-QAM, but also cross QAM, abigger size QAM, or modulations using other BRGC other than QAM.

FIG. 32 shows an example of Bit mapping of Modified-QAM where 256-QAM ismodified using BRGC. FIG. 32 a and FIG. 32 b show mapping of MostSignificant Bits (MSB). Points designated as filled circles representmappings of ones and points designated as blank circles representmappings of zeros. In a same manner, each bit is mapped as shown infigures from (a) through (h) in FIG. 32, until Least SignificantBits(LSB) are mapped. As shown in FIG. 32, Modified-QAM can enable bitdecision using only I or Q axes as conventional QAM, except for a bitwhich is next to MSB (FIG. 32 c and FIG. 32 d). By using thesecharacteristics, a simple receiver can be made by partially modifying areceiver for QAM. An efficient receiver can be implemented by checkingboth I and Q values only when determining bit next to MSB and bycalculating only 1 or Q for the rest of bits. This method can be appliedto Approximate LLR, Exact LLR, or Hard decision.

By using the Modified-QAM or MQAM, which uses the characteristics ofabove BRGC, Non-uniform constellation or NU-MQAM can be made. In theabove equation where Gaussian CDF is used, Pj can be modified to fitMQAM. Just like QAM, in MQAM, two PAMs having I axis and Q axis can beconsidered. However, unlike QAM where a number of points correspondingto a value of each PAM axis are identical, the number of points changesin MQAM. If a number of points that corresponds to jth value of PAM isdefined as nj in a MQAM where a total of M constellation points exist,then Pj can be defined as follows:

$\begin{matrix}{{{\int_{- \infty}^{X_{j}}{\frac{1}{\sqrt{2\pi}}^{- \frac{X^{2}}{2}}\ {x}}} = P_{j}}{{P_{j} = \frac{{\sum\limits_{i = 0}^{i = {j - 1}}n_{i}} + \frac{n_{j}}{2N}}{M}},{n_{0} = 0}}} & \left( {{Eq}.\mspace{14mu} 2} \right)\end{matrix}$

By using the newly defined Pj, MQAM can be transformed into Non-uniformconstellation. Pj can be defined as follows for the example of 256-MQAM.

$P_{j} \in \left\{ {\frac{2.5}{256},\frac{10}{256},\frac{22}{256},\frac{36}{256},\frac{51}{256},\frac{67}{256},\frac{84}{256},\frac{102}{256},\frac{119.5}{256},\frac{136.5}{256},\frac{154}{256},\frac{172}{256},\frac{189}{256},\frac{205}{256},\frac{220}{256},\frac{234}{256},\frac{246}{256},\frac{253.5}{256}} \right\}$

FIG. 33 is an example of transformation of MQAM into Non-uniformconstellation. The NU-MQAM made using these methods can retaincharacteristics of MQAM receivers with modified coordinates of each PAM.Thus, an efficient receiver can be implemented. In addition, a morenoise-robust system than the previous NU-QAM can be implemented. For amore efficient broadcasting transmission system, hybridizing MQAM andNU-MQAM is possible. In other words, a more noise-robust system can beimplemented by using MQAM for an environment where an error correctioncode with high code rate is used and by using NU-MQAM otherwise. Forsuch a case, a transmitter can let a receiver have information of coderate of an error correction code currently used and a kind of modulationcurrently used such that the receiver can demodulate according to themodulation currently used.

FIG. 34 shows an example of digital transmission system. Inputs cancomprise a number of MPEG-TS streams or GSE (General StreamEncapsulation) streams. An input processor module 101 can addtransmission parameters to input stream and perform scheduling for aBICM module 102. The BICM module 102 can add redundancy and interleavedata for transmission channel error correction. A frame builder 103 canbuild frames by adding physical layer signaling information and pilots.A modulator 104 can perform modulation on input symbols in efficientmethods. An analog processor 105 can perform various processes forconverting input digital signals into output analog signals.

FIG. 35 shows an example of an input processor. Input MPEG-TS or GSEstream can be transformed by input preprocessor into a total of nstreams which will be independently processed. Each of those streams canbe either a complete TS frame which includes multiple service componentsor a minimum TS frame which includes service component (i.e., video oraudio). In addition, each of those streams can be a GSE stream whichtransmits either multiple services or a single service.

Input interface module 202-1 can allocate a number of input bits equalto the maximum data field capacity of a Baseband (BB) frame. A paddingmay be inserted to complete the LDPC/BCH code block capacity. The inputstream sync module 203-1 can provide a mechanism to regenerate, in thereceiver, the clock of the Transport Stream (or packetized GenericStream), in order to guarantee end-to-end constant bit rates and delay.

In order to allow the Transport Stream recombining without requiringadditional memory in the receiver, the input Transport Streams aredelayed by delay compensators 204-1˜n considering interleavingparameters of the data PLPs in a group and the corresponding common PLP.Null packet deleting modules 205-1˜n can increase transmissionefficiency by removing inserted null packet for a case of VBR (variablebit rate) service. Cyclic Redundancy Check (CRC) encoder modules 206-1˜ncan add CRC parity to increase transmission reliability of BB frame. BBheader inserting modules 207-1˜n can add BB frame header at a beginningportion of BB frame. Information that can be included in BB header isshown in FIG. 36.

A Merger/slicer module 208 can perform BB frame slicing from each PLP,merging BB frames from multiple PLPs, and scheduling each BB framewithin a transmission frame. Therefore, the merger/slicer module 208 canoutput L1 signaling information which relates to allocation of PLP inframe. Lastly, a BB scrambler module 209 can randomize input bitstreamsto minimize correlation between bits within bitstreams. The modules inshadow in FIG. 35 are modules used when transmission system uses asingle PLP, the other modules in FIG. 35 are modules used when thetransmission device uses multiple PLPs.

FIG. 37 shows an example of BICM module. FIG. 37 a shows data path andFIG. 37 b shows L1 path of BICM module. An outer coder module 301 and aninner coder module 303 can add redundancy to input bitstreams for errorcorrection. An outer interleaver module 302 and an inner interleavermodule 304 can interleave bits to prevent burst error. The Outerinterleaver module 302 can be omitted if the BICM is specifically forDVB-C2. A bit demux module 305 can control reliability of each bitoutput from the inner interleaver module 304. A symbol mapper module 306can map input bitstreams into symbol streams. At this time, it ispossible to use any of a conventional QAM, an MQAM which uses theaforementioned BRGC for performance improvement, an NU-QAM which usesNon-uniform modulation, or an NU-MQAM which uses Non-uniform modulationapplied BRGC for performance improvement. To construct a system which ismore robust against noise, combinations of modulations using MQAM and/orNU-MQAM depending on the code rate of the error correction code and theconstellation capacity can be considered. At this time, the Symbolmapper module 306 can use a proper constellation according to the coderate and constellation capacity. FIG. 39 shows an example of suchcombinations.

Case 1 shows an example of using only NU-MQAM at low code rate forsimplified system implementation. Case 2 shows an example of usingoptimized constellation at each code rate. The transmitter can sendinformation about the code rate of the error correction code and theconstellation capacity to the receiver such that the receiver can use anappropriate constellation. FIG. 40 shows another example of cases wherecompatibility between conventional systems is considered. In addition tothe examples, further combinations for optimizing the system arepossible.

The ModCod Header inserting module 307 shown in FIG. 37 can takeAdaptive coding and modulation (ACM)/Variable coding and modulation(VCM) feedback information and add parameter information used in codingand modulation to a FEC block as header. The Modulation type/Coderate(ModCod) header can include the following information:

-   -   FEC type (1 bits) -long or short LDPC    -   Coderate (3 bits)    -   Modulation (3 bits)-up-to 64K QAM    -   PLP identifier (8 bits)

The Symbol interleaver module 308 can perform interleaving in symboldomain to obtain additional interleaving effects. Similar processesperformed on data path can be performed on L1 signaling path but withpossibly different parameters (301-1˜308-1). At this point, ashortened/punctured code module (303-1) can be used for inner code.

FIG. 38 shows an example of LDPC encoding using shortening/puncturing.Shortening process can be performed on input blocks which have less bitsthan a required number of bits for LDPC encoding as many zero bitsrequired for LDPC encoding can be padded (301 c). Zero Padded inputbitstreams can have parity bits through LDPC encoding (302 c). At thistime, for bitstreams that correspond to original bitstreams, zeros canbe removed (303 c) and for parity bitstreams, puncturing (304 c) can beperformed according to code-rates. These processed informationbitstreams and parity bitstreams can be multiplexed into originalsequences and outputted (305 c).

FIG. 41 shows a frame structure which comprises preamble for L1signaling and data symbol for PLP data. It can be seen that preamble anddata symbols are cyclically generated, using one frame as a unit. Datasymbols comprise PLP type 0 which is transmitted using a fixedmodulation/coding and PLP type 1 which is transmitted using a variablemodulation/coding. For PLP type 0, information such as modulation, FECtype, and FEC code rate are transmitted in preamble (see FIG. 42 Frameheader insert 401). For PLP type 1, corresponding information can betransmitted in FEC block header of a data symbol (see FIG. 37 ModCodheader insert 307). By the separation of PLP types, ModCod overhead canbe reduced by 3-4% from a total transmission rate, for PLP type0 whichis transmitted at a fixed bit rate. At a receiver, for fixedmodulation/coding PLP of PLP type 0, Frame header remover r401 shown inFIG. 63 can extract information on Modulation and FEC code rate andprovide the extracted information to a BICM decoding module. Forvariable modulation/coding PLP of PLP type 1, ModCod extracting modules,r307 and r307-1 shown in FIG. 64 can extract and provide the parametersnecessary for BICM decoding.

FIG. 42 shows an example of a frame builder. A frame header insertingmodule 401 can form a frame from input symbol streams and can add frameheader at front of each transmitted frame. The frame header can includethe following information:

 * Number of bonded channels (4 bits)  * Guard interval (2 bits)  * PAPR(2 bits)  * Pilot pattern (2 bits)  * Digital System identification (16bits)  * Frame identification (16 bits)  * Frame length (16 bits) numberof Orthogonal Frequency Division Multiplexing (OFDM) symbols per frame * Superframe length (16 bits) number of frames per superframe  * numberof PLPs (8 bits)  * for each PLP  PLP identification (8 bits)  Channelbonding id (4 bits)  PLP start (9 bits)  PLP type (2 bits) common PLP orothers  PLP payload type (5 bits)  MC type (1 bit) -fixed/variablemodulation & coding  if MC type == fixed modulation & coding  FEC type(1 bits) -long or short LDPC  Coderate (3 bits)  Modulation (3 bits)-up-to 64K QAM  end if;  Number of notch channels (2 bits)  for eachnotch  Notch start (9 bits)  Notch width (9 bits)  end for;  PLP width(9 bits) - max number of FEC blocks of PLP  PLP time interleaving type(2 bits)  end for;  * CRC-32 (32 bits)

Channel bonding environment is assumed for L1 information transmitted inFrame header and data that correspond to each data slice is defined asPLP. Therefore, information such as PLP identifier, channel bondingidentifier, and PLP start address are required for each channel used inbonding. One embodiment of this invention suggests transmitting ModCodfield in FEC frame header if PLP type supports variablemodulation/coding and transmitting ModCod field in Frame header if PLPtype supports fixed modulation/coding to reduce signaling overhead. Inaddition, if a Notch band exists for each PLP, by transmitting the startaddress of the Notch and its width, decoding corresponding carriers atthe receiver can become unnecessary.

FIG. 43 shows an example of Pilot Pattern 5 (PP5) applied in a channelbonding environment. As shown, if SP positions are coincident withpreamble pilot positions, irregular pilot structure can occur.

FIG. 43 a shows an example of pilot inserting module 404 as shown inFIG. 42. As represented in FIG. 43, if a single frequency band (forexample, 8 MHz) is used, the available bandwidth is 7.61 MHz, but ifmultiple frequency bands are bonded, guard bands can be removed, thus,frequency efficiency can increase greatly. FIG. 43 b is an example ofpreamble inserting module 504 as shown in FIG. 51 that is transmitted atthe front part of the frame and even with channel bonding, the preamblehas repetition rate of 7.61 MHz, which is bandwidth of L1 block. This isa structure considering the bandwidth of a tuner which performs initialchannel scanning.

Pilot Patterns exist for both Preamble and Data Symbols. For datasymbol, scattered pilot (SP) patterns can be used. Pilot Pattern 5 (PP5)and Pilot Pattern 7 (PP7) of T2 can be good candidates forfrequency-only interpolation. PP5 has x=12, y=4, z=48 for GI= 1/64 andPP7 has x=24, y=4, z=96 for GI= 1/128. Additional time-interpolation isalso possible for a better channel estimation. Pilot patterns forpreamble can cover all possible pilot positions for initial channelacquisition. In addition, preamble pilot positions should be coincidentwith SP positions and a single pilot pattern for both the preamble andthe SP is desired. Preamble pilots could also be used fortime-interpolation and every preamble could have an identical pilotpattern. These requirements are important for C2 detection in scanningand necessary for frequency offset estimation with scrambling sequencecorrelation. In a channel bonding environment, the coincidence in pilotpositions should also be kept for channel bonding because irregularpilot structure may degrade interpolation performance.

In detail, if a distance z between scattered pilots (SPs) in an OFDMsymbol is 48 and if a distance y between SPs corresponding to a specificSP carrier along the time axis is 4, an effective distance x after timeinterpolation becomes 12. This is when a guard interval (GI) fraction is1/64. If GI fraction is 1/128, x=24, y=4, and z=96 can be used. Ifchannel bonding is used, SP positions can be made coincident withpreamble pilot positions by generating non-continuous points inscattered pilot structure.

At this time, preamble pilot positions can be coincident with every SPpositions of data symbol. When channel bonding is used, data slice wherea service is transmitted, can be determined regardless of 8 MHzbandwidth granularity. However, for reducing overhead for data sliceaddressing, transmission starting from SP position and ending at SPposition can be chosen.

When a receiver receives such SPs, if necessary, channel estimationmodule r501 shown in FIG. 62 can perform time interpolation to obtainpilots shown in dotted lines in FIG. 43 and perform frequencyinterpolation. At this time, for non-continuous points of whichintervals are designated as 32 in FIG. 43, either performinginterpolations on left and right separately or performing interpolationson only one side then performing interpolation on the other side byusing the already interpolated pilot positions of which interval is 12as a reference point can be implemented. At this time, data slice widthcan vary within 7.61 MHz, thus, a receiver can minimize powerconsumption by performing channel estimation and decoding only necessarysubcarriers.

FIG. 44 shows another example of PP5 applied in channel bondingenvironment or a structure of SP for maintaining effective distance x as12 to avoid irregular SP structure shown in FIG. 43 when channel bondingis used. FIG. 44 a is a structure of SP for data symbol and FIG. 44 b isa structure of SP for preamble symbol.

As shown, if SP distance is kept consistent in case of channel bonding,there will be no problem in frequency interpolation but pilot positionsbetween data symbol and preamble may not be coincident. In other words,this structure does not require additional channel estimation for anirregular SP structure, however, SP positions used in channel bondingand preamble pilot positions become different for each channel.

FIG. 45 shows a new SP structure or PP5 to provide a solution to the twoproblems aforementioned in channel bonding environment. Specifically, apilot distance of x=16 can solve those problems. To preserve pilotdensity or to maintain the same overhead, a PP5′ can have x=16, y=3,z=48 for GI= 1/64 and a PP7′ can have x=16, y=6, z=96 for GI= 1/128.Frequency-only interpolation capability can still be maintained. Pilotpositions are depicted in FIG. 45 for comparison with PP5 structure.

FIG. 46 shows an example of a new SP Pattern or PP5 structure in channelbonding environment. As shown in FIG. 46, whether either single channelor channel bonding is used, an effective pilot distance x=16 can beprovided. In addition, because SP positions can be made coincident withpreamble pilot positions, channel estimation deterioration caused by SPirregularity or non-coincident SP positions can be avoided. In otherwords, no irregular SP position exists for freq-interpolator andcoincidence between preamble and SP positions is provided.

Consequently, the proposed new SP patterns can be advantageous in thatsingle SP pattern can be used for both single and bonded channel; noirregular pilot structure can be caused, thus a good channel estimationis possible; both preamble and SP pilot positions can be keptcoincident; pilot density can be kept the same as for PP5 and PP7respectively; and Frequency-only interpolation capability can also bepreserved.

In addition, the preamble structure can meet the requirements such aspreamble pilot positions should cover all possible SP positions forinitial channel acquisition; maximum number of carriers should be 3409(7.61 MHz) for initial scanning; exactly same pilot patterns andscrambling sequence should be used for C2 detection; and nodetection-specific preamble like P1 in T2 is required.

In terms of relation with frame structure, data slice positiongranularity may be modified to 16 carriers rather than 12, thus, lessposition addressing overhead can occur and no other problem regardingdata slice condition, Null slot condition etc can be expected.

Therefore, at channel estimation module r501 of FIG. 62, pilots in everypreamble can be used when time interpolation of SP of data symbol isperformed. Therefore, channel acquisition and channel estimation at theframe boundaries can be improved.

Now, regarding requirements related to the preamble and the pilotstructure, there is consensus in that positions of preamble pilots andSPs should coincide regardless of channel bonding; the number of totalcarriers in L1 block should be dividable by pilot distance to avoidirregular structure at band edge; L1 blocks should be repeated infrequency domain; and L1 blocks should always be decodable in arbitrarytuner window position. Additional requirements would be that pilotpositions and patterns should be repeated by period of 8 MHz; correctcarrier frequency offset should be estimated without channel bondingknowledge; and L1 decoding (re-ordering) is impossible before thefrequency offset is compensated.

FIG. 47 shows a relationship between data symbol and preamble whenpreamble structures as shown in FIG. 52 and FIG. 53 are used. L1 blockcan be repeated by period of 6 MHz. For L1 decoding, both frequencyoffset and Preamble shift pattern should be found. L1 decoding is notpossible in arbitrary tuner position without channel bonding informationand a receiver cannot differentiate between preamble shift value andfrequency offset.

Thus, a receiver, specifically for Frame header remover r401 shown inFIG. 63 to perform L1 signal decoding, channel bonding structure needsto be obtained. Because preamble shift amount expected at two verticallyshadowed regions in FIG. 47 is known, time/freq synchronizing moduler505 in FIG. 62 can estimate carrier frequency offset. Based on theestimation, L1 signaling path (r308-1˜r301-1) in FIG. 64 can decode L1.

FIG. 48 shows a relationship between data symbol and preamble when thepreamble structure as shown in FIG. 55 is used. L1 block can be repeatedby period of 8 MHz. For L1 decoding, only frequency offset needs to befound and channel bonding knowledge may not be required. Frequencyoffset can be easily estimated by using known Pseudo Random BinarySequence (PRBS) sequence. As shown in FIG. 48, preamble and data symbolsare aligned, thus, additional sync search can become unnecessary.Therefore, for a receiver, specifically for the Frame header removermodule r401 shown in FIG. 63, it is possible that only correlation peakwith pilot scrambling sequence needs to be obtained to perform L1 signaldecoding. The time/freq synchronizing module r505 in FIG. 62 canestimate carrier frequency offset from peak position.

FIG. 49 shows an example of cable channel delay profile.

From the point of view of pilot design, current GI already over-protectsdelay spread of cable channel. In the worst case, redesigning thechannel model can be an option. To repeat the pattern exactly every 8MHz, the pilot distance should be a divisor of 3584 carriers (z=32 or56). A pilot density of z=32 can increase pilot overhead, thus, z=56 canbe chosen. Slightly less delay coverage may not be an important in cablechannel. For example, it can be 8 μs for PP5′ and 4 μs for PP7′ comparedto 9.3 μs (PP5) and 4.7 μs (PP7). Meaningful delays can be covered byboth pilot patterns even in a worst case. For preamble pilot position,no more than all SP positions in data symbol are necessary.

If the −40 dB delay path can be ignored, actual delay spread can become2.5 us, 1/64 GI=7 μs, or 1/128 GI=3.5 μs. This shows that pilot distanceparameter, z=56 can be a good enough value. In addition, z=56 can be aconvenient value for structuring pilot pattern that enables preamblestructure shown in FIG. 48.

FIG. 50 shows scattered pilot structure that uses z=56 and z=112 whichis constructed at pilot inserting module 404 in FIG. 42. PP5′ (x=14,y=4, z=56) and PP7′ (x=28, y=4, z=112) are proposed. Edge carriers couldbe inserted for closing edge.

As shown in FIG. 50, pilots are aligned at 8 MHz from each edge of theband, every pilot position and pilot structure can be repeated every 8MHz. Thus, this structure can support the preamble structure shown inFIG. 48. In addition, a common pilot structure between preamble and datasymbols can be used. Therefore, channel estimation module r501 in FIG.62 can perform channel estimation using interpolation on preamble anddata symbols because no irregular pilot pattern can occur, regardless ofwindow position which is decided by data slice locations. At this time,using only frequency interpolation can be enough to compensate channeldistortion from delay spread. If time interpolation is performedadditionally, more accurate channel estimation can be performed.

Consequently, in the new proposed pilot pattern, pilot position andpattern can be repeated based on a period of 8 MHz. A single pilotpattern can be used for both preamble and data symbols. L1 decoding canalways be possible without channel bonding knowledge. In addition, theproposed pilot pattern may not affect commonality with T2 because thesame pilot strategy of scattered pilot pattern can be used; T2 alreadyuses 8 different pilot patterns; and no significant receiver complexitycan be increased by modified pilot patterns. For a pilot scramblingsequence, the period of PRBS can be 2047 (m-sequence); PRBS generationcan be reset every 8 MHz, of which the period is 3584; pilot repetitionrate of 56 can be also co-prime with 2047; and no PAPR issue can beexpected.

FIG. 51 shows an example of a modulator based on OFDM. Input symbolstreams can be transformed into time domain by IFFT module 501. Ifnecessary, peak-to-average power ratio (PAPR) can be reduced at PAPRreducing module 502. For PAPR methods, Active constellation extension(ACE) or tone reservation can be used. GI inserting module 503 can copya last part of effective OFDM symbol to fill guard interval in a form ofcyclic prefix.

Preamble inserting module 504 can insert preamble at the front of eachtransmitted frame such that a receiver can detect digital signal, frameand acquire time/freq offset acquisition. At this time, the preamblesignal can perform physical layer signaling such as FFT size (3 bits)and Guard interval size (3 bits). The Preamble inserting module 504 canbe omitted if the modulator is specifically for DVB-C2.

FIG. 52 shows an example of a preamble structure for channel bonding,generated at preamble inserting module 504 in FIG. 51. One complete L1block should be “always decodable” in any arbitrary 7.61 MHz tuningwindow position and no loss of L1 signaling regardless of tuner windowposition should occur. As shown, L1 blocks can be repeated in frequencydomain by period of 6 MHz. Data symbol can be channel bonded for every 8MHz. If, for L1 decoding, a receiver uses a tuner such as the tuner r603represented in FIG. 61 which uses a bandwidth of 7.61 MHz, the Frameheader remover r401 in FIG. 63 needs to rearrange the received cyclicshifted L1 block (FIG. 53) to its original form. This rearrangement ispossible because L1 block is repeated for every 6 MHz block. FIG. 53 acan be reordered into FIG. 53 b.

FIG. 54 shows a process for designing a more optimized preamble. Thepreamble structure of FIG. 52 uses only 6 MHz of total tuner bandwidth7.61 MHz for L1 decoding. In terms of spectrum efficiency, tunerbandwidth of 7.61 MHz is not fully utilized. Therefore, there can befurther optimization in spectrum efficiency.

FIG. 55 shows another example of preamble structure or preamble symbolsstructure for full spectrum efficiency, generated at Frame HeaderInserting module 401 in FIG. 42. Just like data symbol, L1 blocks can berepeated in frequency domain by period of 8 MHz. One complete L1 blockis still “always decodable” in any arbitrary 7.61 MHz tuning windowposition. After tuning, the 7.61 MHz data can be regarded as a virtuallypunctured code. Having exactly the same bandwidth for both the preambleand data symbols and exactly the same pilot structure for both thepreamble and data symbols can maximize spectrum efficiency. Otherfeatures such as cyclic shifted property and not sending L1 block incase of no data slice can be kept unchanged. In other words, thebandwidth of preamble symbols can be identical with the bandwidth ofdata symbols or, as shown in FIG. 57, the bandwidth of the preamblesymbols can be the bandwidth of the tuner (here, it's 7.61 MHz). Thetuner bandwidth can be defined as a bandwidth that corresponds to anumber of total active carriers when a single channel is used. That is,the bandwidth of the preamble symbol can correspond to the number oftotal active carriers (here, it's 7.61 MHz).

FIG. 56 shows a virtually punctured code. The 7.61 MHz data among the 8MHz L1 block can be considered as punctured coded. When a tuner r603shown in FIG. 61 uses 7.61 MHz bandwidth for L1 decoding, Frame headerremover r401 in FIG. 63 needs to rearrange received, cyclic shifted L1block into original form as shown in FIG. 56. At this time, L1 decodingis performed using the entire bandwidth of the tuner. Once the L1 blockis rearranged, a spectrum of the rearranged L1 block can have a blankregion within the spectrum as shown in upper right side of FIG. 56because an original size of L 1 block is 8 MHz bandwidth.

Once the blank region is zero padded, either after deinterleaving insymbol domain by the freq. deinterleaver r403 in FIG. 63 or by thesymbol deinterleaver r308-1 in FIG. 64 or after deinterleaving in bitdomain by the symbol demapper r306-1, bit mux r305-1, and innerdeinterleaver r304-1 in FIG. 64, the block can have a form which appearsto be punctured as shown in lower right side of FIG. 56.

This L1 block can be decoded at the punctured/shortened decode moduler303-1 in FIG. 64. By using these preamble structure, the entire tunerbandwidth can be utilized, thus spectrum efficiency and coding gain canbe increased. In addition, an identical bandwidth and pilot structurecan be used for the preamble and data symbols.

In addition, if the preamble bandwidth or the preamble symbols bandwidthis set as a tuner bandwidth as shown in FIG. 58, (it's 7.61 MHz in theexample), a complete L1 block can be obtained after rearrangement evenwithout puncturing. In other words, for a frame having preamble symbols,wherein the preamble symbols have at least one layer 1 (L1) block, itcan be said, the L1 block has 3408 active subcarriers and the 3408active subcarriers correspond to 7.61 MHz of 8MHz Radio Frequency (RF)band.

Thus, spectrum efficiency and L1 decoding performance can be maximized.In other words, at a receiver, decoding can be performed atpunctured/shortened decode module r303-1 in FIG. 64, after performingonly deinterleaving in the symbol domain.

Consequently, the proposed new preamble structure can be advantageous inthat it's fully compatible with previously used preamble except that thebandwidth is different; L1 blocks are repeated by period of 8 MHz; L1block can be always decodable regardless of tuner window position; Fulltuner bandwidth can be used for L1 decoding; maximum spectrum efficiencycan guarantee more coding gain; incomplete L1 block can be considered aspunctured coded; simple and same pilot structure can be used for bothpreamble and data; and identical bandwidth can be used for both preambleand data.

FIG. 59 shows an example of an analog processor. A DAC module 601 canconvert digital signal input into analog signal. After transmissionfrequency bandwidth is up-converted (602) and analog filtered (603)signal can be transmitted.

FIG. 60 shows an example of a digital receiver system. Received signalis converted into digital signal at an analog process module r105. Ademodulator r104 can convert the signal into data in frequency domain. Aframe parser r103 can remove pilots and headers and enable selection ofservice information that needs to be decoded. A BICM demodulator r102can correct errors in the transmission channel. An output processor r101can restore the originally transmitted service stream and timinginformation.

FIG. 61 shows an example of analog processor used at the receiver. ATuner/AGC module r603 can select desired frequency bandwidth fromreceived signal. A down converting module r602 can restore baseband. AnADC module r601 can convert analog signal into digital signal.

FIG. 62 shows an example of demodulator. A frame detecting module r506can detect the preamble, check if a corresponding digital signal exists,and detect a start of a frame. A time/freq synchronizing module r505 canperform synchronization in time and frequency domains. At this time, fortime domain synchronization, a guard interval correlation can be used.For frequency domain synchronization, correlation can be used or offsetcan be estimated from phase information of a subcarrier that istransmitted in the frequency domain. A preamble removing module r504 canremove preamble from the front of detected frame. A GI removing moduler503 can remove guard interval. A FFT module r501 can transform signalin the time domain into signal in the frequency domain. A channelestimation/equalization module r501 can compensate errors by estimatingdistortion in transmission channel using pilot symbol. The Preambleremoving module r504 can be omitted if the demodulator is specificallyfor DVB-C2.

FIG. 63 shows an example of frame parser. A pilot removing module r404can remove pilot symbol. A freq deinterleaving module r403 can performdeinterleaving in the frequency domain. An OFDM symbol merger r402 canrestore data frame from symbol streams transmitted in OFDM symbols. Aframe header removing module r401 can extract physical layer signalingfrom header of each transmitted frame and remove header. Extractedinformation can be used as parameters for following processes in thereceiver.

FIG. 64 shows an example of a BICM demodulator. FIG. 64 a shows a datapath and FIG. 64 b shows a L1 signaling path. A symbol deinterleaverr308 can perform deinterleaving in the symbol domain. A ModCod extractr307 can extract ModCod parameters from front of each BB frame and makethe parameters available for following adaptive/variable demodulationand decoding processes. A Symbol demapper r306 can demap input symbolstreams into bit Log-Likelyhood Ratio (LLR) streams. The Output bit LLRstreams can be calculated by using a constellation used in a Symbolmapper 306 of the transmitter as reference point. At this point, whenthe aforementioned MQAM or NU-MQAM is used, by calculating both I axisand Q axis when calculating bit nearest from MSB and by calculatingeither I axis or Q axis when calculating the rest bits, an efficientsymbol demapper can be implemented. This method can be applied to, forexample, Approximate LLR, Exact LLR, or Hard decision.

When an optimized constellation according to constellation capacity andcode rate of error correction code at the Symbol mapper 306 of thetransmitter is used, the Symbol demapper r306 of the receiver can obtaina constellation using the code rate and constellation capacityinformation transmitted from the transmitter. The bit mux r305 of thereceiver can perform an inverse function of the bit demux 305 of thetransmitter. The Inner deinterleaver r304 and outer deinterleaver r302of the receiver can perform inverse functions of the inner interleaver304 and outer interleaver 302 of the transmitter, respectively to getthe bitstream in its original sequence. The outer deinterleaver r302 canbe omitted if the BICM demodulator is specifically for DVB-C2.

The inner decoder r303 and outer decoder r301 of the receiver canperform corresponding decoding processes to the inner coder 303 andouter code 301 of the transmitter, respectively, to correct errors inthe transmission channel. Similar processes performed on data path canbe performed on L1 signaling path, but with different parameters(r308-1˜r301-1). At this point, as explained in the preamble part, ashortened/punctured code module r303-1 can be used for L1 signaldecoding.

FIG. 65 shows an example of LDPC decoding using shortening/puncturing. Ademux r301 a can separately output information part and parity part ofsystematic code from input bit streams. For the information part, a zeropadding (r302 a) can be performed according to a number of input bitstreams of LDPC decoder, for the parity part, input bit streams for(r303 a) the LDPC decoder can be generated by depuncturing puncturedpart. LDPC decoding (r304 a) can be performed on generated bit streams,zeros in information part can be removed and output (r305 a).

FIG. 66 shows an example of output processor. A BB descrambler r209 canrestore scrambled (209) bit streams at the transmitter. A Splitter r208can restore BB frames that correspond to multiple PLP that aremultiplexed and transmitted from the transmitter according to PLP path.For each PLP path, a BB header remover r207-1˜n can remove the headerthat is transmitted at the front of the BB frame. A CRC decoder r206-1˜ncan perform CRC decoding and make reliable BB frames available forselection. A Null packet inserting modules r205-1˜n can restore nullpackets which were removed for higher transmission efficiency in theiroriginal location. A Delay recovering modules r204-1˜n can restore adelay that exists between each PLP path.

An output clock recovering modules r203-1˜n can restore the originaltiming of the service stream from timing information transmitted fromthe input stream synchronization modules 203-1˜n. An output interfacemodules r202-1˜n can restore data in TS/GS packet from input bit streamsthat are sliced in BB frame. An output postprocess modules r201-1˜n canrestore multiple TS/GS streams into a complete TS/GS stream, ifnecessary. The shaded blocks shown in FIG. 66 represent modules that canbe used when a single PLP is processed at a time and the rest of theblocks represent modules that can be used when multiple PLPs areprocessed at the same time.

Preamble pilot patterns were carefully designed to avoid PAPR increase,thus, whether L1 repetition rate may increase PAPR needs to beconsidered. The number of L1 information bits varies dynamicallyaccording to the channel bonding, the number of PLPs, etc. In detail, itis necessary to consider things such as fixed L1 block size mayintroduce unnecessary overhead; L1 signaling should be protected morestrongly than data symbols; and time interleaving of L1 block canimprove robustness over channel impairment such as impulsive noise need.

For a L1 block repetition rate of 8 MHz, as shown in FIG. 67, fullspectrum efficiency (26.8% BW increase) is exhibited with virtualpuncturing but the PAPR may be increased since L1 bandwidth is the sameas that of the data symbols. For the repetition rate of 8 MHz, 4K-FFTDVB-T2 frequency interleaving can be used for commonality and the samepattern can repeat itself at a 8 MHz period after interleaving.

For a L1 block repetition rate of 6 MHz, as shown in FIG. 68, reducedspectrum efficiency can be exhibited with no virtual puncturing. Asimilar problem of PAPR as for the 8 MHz case can occur since the L1 anddata symbol bandwidths share LCM=24 MHz. For the repetition rate of 6MHz, 4K-FFT DVB-T2 frequency interleaving can be used for commonalityand the same pattern can repeat itself at a period of 24 MHz afterinterleaving.

FIG. 69 shows a new L1 block repetition rate of 7.61 MHz or full tunerbandwidth. A full spectrum efficiency (26.8% BW increase) can beobtained with no virtual puncturing. There can be no PAPR issue since L1and data symbol bandwidths share LCM≈1704 MHz. For the repetition rateof 7.61 MHz, 4K-FFT DVB-T2 frequency interleaving can be used forcommonality and the same pattern can repeat itself by period of about1704 MHz after interleaving.

FIG. 70 is an example of L1 signaling which is transmitted in the frameheader. Each information in L1 signaling can be transmitted to thereceiver and can be used as a decoding parameter. Especially, theinformation can be used in L1 signal path shown in FIG. 64 and PLPs canbe transmitted in each data slice. An increased robustness for each PLPcan be obtained.

FIG. 72 is an example of a symbol interleaver 308-1 as shown in L1signaling path in

FIG. 37 and also can be an example of its corresponding symboldeinterleaver r308-1 as shown in L1 signaling path in FIG. 64. Blockswith tilted lines represent L1 blocks and solid blocks represent datacarriers. L1 blocks can be transmitted not only within a singlepreamble, but also can be transmitted within multiple OFDM blocks.Depending on a size of L1 block, the size of the interleaving block canvary. In other words, num_L1_sym and L1 span can be different from eachother. To minimize unnecessary overhead, data can be transmitted withinthe rest of the carriers of the OFDM symbols where the L1 block istransmitted. At this point, full spectrum efficiency can be guaranteedbecause the repeating cycle of L1 block is still a full tuner bandwidth.In FIG. 72, the numbers in blocks with tilted lines represent the bitorder within a single LDPC block.

Consequently, when bits are written in an interleaving memory in the rowdirection according to a symbol index as shown in FIG. 72 and read inthe column direction according to a carrier index, a block interleavingeffect can be obtained. In other words, one LDPC block can beinterleaved in the time domain and the frequency domain and then can betransmitted. Num_L1_sym can be a predetermined value, for example, anumber between 2˜4 can be set as a number of OFDM symbols. At thispoint, to increase the granularity of the L1 block size, apunctured/shortened LDPC code having a minimum length of the codewordcan be used for L1 protection.

FIG. 73 is an example of an L1 block transmission. FIG. 73 illustratesFIG. 72 in frame domain. As shown on FIG. 73 a, L1 blocks can bespanning in full tuner bandwidth or as shown on FIG. 73 b, L1 blocks canbe partially spanned and the rest of the carriers can be used for datacarrier. In either case, it can be seen that the repetition rate of L1block can be identical to a full tuner bandwidth. In addition, for OFDMsymbols which uses L1 signaling including preamble, only symbolinterleaving can be performed while not allowing data transmission inthat OFDM symbols. Consequently, for OFDM symbol used for L1 signaling,a receiver can decode L1 by performing deinterleaving without datadecoding. At this point, the L1 block can transmit L1 signaling ofcurrent frame or L1 signaling of a subsequent frame. At the receiverside, L1 parameters decoded from L1 signaling decoding path shown inFIG. 64 can be used for decoding process for data path from frame parserof subsequent frame.

In summary, at a transmitter, interleaving blocks of L1 region can beperformed by writing blocks to a memory in a row direction and readingthe written blocks from the memory in a column direction. At a receiver,deinterleaving blocks of L1 region can be performed by writing blocks toa memory in a column direction and reading the written blocks from thememory in a row direction. The reading and writing directions oftransmitter and receiver can be interchanged.

When simulation is performed with assumptions such as CR=½ for L1protection and for T2 commonality; 16-QAM symbol mapping; pilot densityof 6 in the Preamble; number of short LDPC implies required amount ofpuncturing/shortening are made, results or conclusions such as onlypreamble for L1 transmission may not be sufficient; the number of OFDMsymbols depends on the amount of L1 block size; shortest LDPC codeword(e.g. 192 bits information) among shortened/punctured code may be usedfor flexibility and fine granularity; and Padding may be added ifrequired with negligible overhead, can be obtained. The result issummarized in FIG. 71.

Consequently, for a L1 block repetition rate, full tuner bandwidth withno virtual puncturing can be a good solution and still no PAPR issue canarise with full spectrum efficiency. For L1 signaling, efficientsignaling structure can allow maximum configuration in an environment of8 channels bonding, 32 notches, 256 data slices, and 256 PLPs. For L1block structure, flexible L1 signaling can be implemented according toL1 block size. Time interleaving can be performed for better robustnessfor T2 commonality. Less overhead can allow data transmission inpreamble.

Block interleaving of L1 block can be performed for better robustness.The interleaving can be performed with fixed pre-defined number of L1symbols (num_L1_sym) and a number of carriers spanned by L1 as aparameter (L1_span). The same technique is used for P2 preambleinterleaving in DVB-T2.

L1 block of variable size can be used. Size can be adaptable to theamount of L1 signaling bits, resulting in a reduced overhead. Fullspectrum efficiency can be obtained with no PAPR issue. Less than 7.61MHz repetition can mean that more redundancy can be sent but unused. NoPAPR issue can arise because of 7.61 MHz repetition rate for L1 block.

FIG. 74 is another example of L1 signaling transmitted within a frameheader. This FIG. 74 is different from FIG. 70 in that the L1_span fieldhaving 12 bits it is divided into two fields. In other words, theL1_span field is divided into a L1_column having 9 bits and a L1_rowhaving 3 bits. The L1_column represents the carrier index that L1 spans.Because data slice starts and ends at every 12 carriers, which is thepilot density, the 12 bits of overhead can be reduced by 3 bits to reach9 bits.

L1_row represents the number of OFDM symbols where L1 is spanning whentime interleaving is applied. Consequently, time interleaving can beperformed within an area of L1_columns multiplied by L1_rows.Alternatively, a total size of L1 blocks can be transmitted such thatL1_span shown in FIG. 70 can be used when time interleaving is notperformed. For such a case, L1 block size is 11,776×2 bits in theexample, thus 15 bits is enough. Consequently, the L1_span field can bemade up of 15 bits.

FIG. 75 is an example of frequency or time interleaving/deinterleaving.The FIG. 75 shows a part of a whole transmission frame. The FIG. 75 alsoshows bonding of multiple 8 MHz bandwidths. A frame can consist of apreamble which transmits L1 blocks and a data symbol which transmitsdata. The different kinds of data symbols represent data slices fordifferent services. As shown in FIG. 75, the preamble transmits L1blocks for every 7.61 MHz.

For the preamble, frequency or time interleaving is performed within L1blocks and not performed between L1 blocks. That is, for the preamble,it can be said that interleaving is performed at L1 block level. Thisallows decoding the L1 blocks by transmitting L1 blocks within a tunerwindow bandwidth even when the tuner window has moved to a randomlocation within a channel bonding system.

For decoding data symbol at a random tuner window bandwidth,interleaving between data slices should not occur. That is, for dataslices, it can be said that interleaving is performed at data slicelevel. Consequently, frequency interleaving and time interleaving shouldbe performed within a data slice. Therefore, a symbol interleaver 308 ina data path of a BICM module of transmitter as shown in FIG. 37 canperform symbol interleaving for each data slice. A symbol interleaver308-1 in an L1 signal path can perform symbol interleaving for each L1block.

A frequency interleaver 403 shown in FIG. 42 needs to performinterleaving on the preamble and data symbols separately. Specifically,for the preamble, frequency interleaving can be performed for each L1block and for data symbol, frequency interleaving can be performed foreach data slice. At this point, time interleaving in data path or L1signal path may not be performed considering low latency mode.

Using the suggested methods and devices, among others advantages it ispossible to implement an efficient digital transmitter, receiver andstructure of physical layer signaling.

By transmitting ModCod information in each BB frame header that isnecessary for ACM/VCM and transmitting the rest of the physical layersignaling in a frame header, signaling overhead can be minimized.

Modified QAM for a more energy efficient transmission or a morenoise-robust digital broadcasting system can be implemented. The systemcan include transmitter and receiver for each example disclosed and thecombinations thereof.

An Improved Non-uniform QAM for a more energy efficient transmission ora more noise-robust digital broadcasting system can be implemented. Amethod of using code rate of error correction code of NU-MQAM and MQAMis also described. The system can include transmitter and receiver foreach example disclosed and the combinations thereof.

The suggested L1 signaling method can reduce overhead by 3-4% byminimizing signaling overhead during channel bonding.

It will be apparent to those skilled in the art that variousmodifications and variations can be made in the present inventionwithout departing from the invention.

1-15. (canceled)
 16. A method for transmitting broadcast signals, themethod comprising: processing input streams in order to output PLP(Physical Layer Pipe) data corresponding to PLPs; encoding preamble datacarrying L1 (Layer 1) signaling information related to allocation of thePLPs, the L1 signaling information including identification informationfor identifying the PLP data and data unit information for identifying adata unit to access the PLP data; encoding the PLP data by an LDPCscheme; QAM (Quadrature Amplitude Modulation) mapping the encoded PLPdata; time interleaving the mapped PLP data; building signal frames,wherein at least one signal frame includes the encoded preamble data anddata symbols including the time interleaved PLP data; modulating thebuilt signal frame by an Orthogonal Frequency Division Multiplexing(OFDM) method; and transmitting the modulated signal frame.
 17. Themethod of claim 16, wherein a length of the data unit information is 8bits.
 18. The method of claim 16, wherein the L1 signaling informationfurther includes type information indicating a type of each PLP.
 19. Anapparatus for transmitting broadcast signals, the apparatus comprising:a processor configured to process input streams in order to output PLP(Physical Layer Pipe) data corresponding to PLPs; a preamble encoderconfigured to encode preamble data carrying L1 (Layer 1) signalinginformation related to allocation of the PLPs, the L1 signalinginformation including identification information for identifying the PLPdata and data unit information for identifying a data unit to access thePLP data; an encoder configured to encode the PLP data by an LDPCscheme; a mapper configured to QAM (Quadrature Amplitude Modulation) mapthe encoded PLP data; a time interleaver configured to time interleavethe mapped PLP data; a builder configured to build signal frames,wherein at least one signal frame includes the encoded preamble data anddata symbols including the time interleaved PLP data; a modulatorconfigured to modulate the built signal frame by an Orthogonal FrequencyDivision Multiplexing (OFDM) method; and a transmitter configured totransmit the modulated signal frame.
 20. The apparatus of claim 19,wherein a length of the data unit information is 8 bits.
 21. Theapparatus of claim 19, wherein the L1 signaling information furtherincludes type information indicating a type of each PLP.
 22. A methodfor processing broadcast signals, the method comprising: receiving thebroadcast signals; demodulating the received broadcast signals by anOrthogonal Frequency Division Multiplexing (OFDM) method; obtainingsignal frame from the demodulated broadcast signals, wherein at leastone signal frames includes preamble data and data symbols including PLP(Physical Layer Pipe) data corresponding to PLPs, wherein the preambledata carries L1 (Layer 1) signaling information related to allocation ofthe PLPs, the L1 signaling information including identificationinformation for identifying the PLP data and data unit information foridentifying a data unit to access the PLP data; time deinterleaving thePLP data; QAM (Quadrature Amplitude Modulation) demapping the timedeinterleaved PLP data; decoding the demapped PLP data; decoding thepreamble data; and output processing the decoded PLP data.
 23. Themethod of claim 22, wherein a length of the data unit information is 8bits.
 24. The method of claim 22, wherein the L1 signaling informationfurther includes type information indicating a type of each PLP.
 25. Anapparatus for processing broadcast signals, the apparatus comprising: areceiver configured to receive the broadcast signals; a demodulatorconfigured to demodulate the received broadcast signals by an OrthogonalFrequency Division Multiplexing (OFDM) method; a frame parser configuredto obtain signal frames from the demodulated broadcast signals, whereinat least one signal frame includes preamble data and data symbolsincluding PLP (Physical Layer Pipe) data corresponding to PLPs, whereinthe preamble data carries L1 (Layer 1) signaling information related toallocation of the PLPs, the L1 signaling information includingidentification information for identifying the PLP data and data unitinformation for identifying a data unit to access the PLP data; a timedeinterleaver configured to time deinterleave the PLP data; a demapperconfigured to QAM (Quadrature Amplitude Modulation) demap the timedeinterleaved PLP data; a first decoder configured to decode thedemapped PLP data; a second decoder configured to decode the preambledata; and a processor configured to process the decoded PLP data. 26.The apparatus of claim 25, wherein a length of the data unit informationis 8 bits.
 27. The apparatus of claim 25, wherein the L1 signalinginformation further includes type information indicating a type of eachPLP.